1. Field
Embodiments of the invention relate to electronic systems, and more particularly, to radio frequency (RF) transmitters in electronic systems.
2. Description of the Related Technology
Certain electronic systems employ a wireless transmitter, for example, a radio frequency transmitter. One challenge of designing high-performance radio transmitters is achieving spectral purity with minimum noise or distortion. Ideally, a radio frequency transmitter produces a desired signal and no noise or distortion.
In practice, however, transmitters also produce and transmit noise and spurious signals in addition to the desired signal. The transmitted noise can degrade either network or device performance in a variety of ways. For example, in a frequency division duplex (FDD) system, transmitter noise that falls on a receive channel can desensitize a receiver. Furthermore, in any system, transmitter noise can fall in the receive channel of other devices and interfere with their performance. Under some frequency licensing schemes (for example, under rules such as 37 C.F.R. §90.18), there may be strict limits regarding how much out-of-band noise may be transmitted. Conversely, in-band noise restrictions may be relatively lenient. Therefore, there is a need for providing a wireless transmitter with a scheme to minimize undesired noise.
Overview of Radio Frequency Transmitter
FIG. 1 is a block diagram illustrating a conventional linear radio frequency transmitter 100. The illustrated transmitter 100 includes a transmission path which includes a digital modulator 110 which has in-phase (I) and quadrature (Q) outputs, first and second digital-to-analog converters (DACs) 120, 140, first and second low pass filters 125, 145, first and second mixers 135, 155, first and second local oscillators 130, 150, a summing unit 160, a band-pass filter (BPF) 165, an amplifier 170, and a front-end module (FEM) 175.
The digital modulator 110 serves to convert a data input into a digital symbol using a mapping such as N-QAM, for example.
The first DAC 120 serves to receive the in-phase component of the modulated signal, and converts it into a first analog signal. The second DAC 140 serves to receive the quadrature-phase component of the modulated signal, and converts it into a second analog signal.
The first low-pass filter 125 serves to select a desired frequency range and block undesired frequencies in the first analog signal along the in-phase path. The first low-pass filter 125 generates a first filtered analog signal. The first low-pass filter 125 can act as a reconstruction filter to remove images introduced by the first DAC 120. The second low-pass filter 145 serves to select a desired frequency range and block undesired frequencies in the second analog signal along the quadrature-phase path. The second low-pass filter 145 generates a second filtered analog signal. Similarly to the first low-pass filter 125, the second low-pass filter 145 can act as a reconstruction filter to remove images introduced by the second DAC 140.
The first local oscillator (ILO) 130 is an in-phase local oscillator that serves to generate an in-phase RF carrier frequency for modulation by the first filtered analog signal. The second local oscillator (QLO) 150 is a quadrature-phase local oscillator that serves to generate a quadrature-phase RF carrier frequency for modulation by the second filtered analog signal. The ILO 130 and the QLO 150 can be implemented as a single unit that outputs two local frequency signals with a phase difference of about 90 degrees from each other.
The first mixer 135 serves to modulate the in-phase RF carrier frequency, generated by the ILO 130, by the first filtered analog signal received from the LPF 125. The first mixer 135 generates a first mixed signal. The second mixer 155 serves to modulate the quadrature-phase RF carrier frequency, generated by the QLO 150, by the second filtered analog signal received from the LPF 145. The second mixer 155 generates a second mixed signal.
The summing unit 160 combines the first mixed signal received from the first mixer 135 with the second mixed signal received from the second mixer 155. The summing unit 160 can combine the first and second mixed signals by summing their signals, thereby combining the in-phase and quadrature-phase components into a combined signal.
The band-pass filter (BPF) 165 serves to allow a selected frequency range to pass while rejecting frequencies above and below a desired range. The BPF 165 filters the combined signal received from the summing unit 160 and generates a combined filtered signal.
The amplifier 170 serves to increase the power of the combined filtered signal. The amplifier 170 amplifies the combined filtered signal received from the BPF 165 and generates an amplified, combined, and filtered signal.
The front-end module (FEM) 175 serves to prepare the amplified, combined, and filtered signal for transmission at the antenna 180. The FEM 175 can include a duplex filter or a transmit/receive switch.
The antenna 180 is configured to transmit a wireless signal via a wireless medium, such as air. The antenna 180 transmits the modulated RF signal received from the FEM 175. The antenna 180 can be any suitable antenna for wireless signal reception and transmission.
As semiconductor processing technology evolves, transmitters are being designed under smaller-scale processes because deep-submicron (for example, 65 nm) CMOS technology provides a number of advantages for RF circuits. For example, a relatively high transition frequency (fT) reduces internal node capacitances and enables inductor-less topologies. Furthermore, much faster switching reduces the noise contribution of large signal circuits, such as local oscillator (LO) dividers and buffers. However, these advantages are typically accompanied by challenges, such as lower supply voltage (for example, 1.2 V) and lower intrinsic transistor gain (gm*r0). As the supply voltage decreases, the traditional design challenges reverse. The relatively fast switching of deep-submicron transistors allows the LO path to contribute less noise and to operate with acceptable power efficiency. On the other hand, it has become more of a challenge to have relatively good noise characteristics in the baseband signal path under a low-voltage supply.
Traditionally, the transmitter 100 operated at a relatively high supply voltage, for example, about +2.7 V. Because the phase noise of local oscillator (LO) paths (for example, the paths starting at the ILO 130 and the QLO 150) can be much higher for a given power efficiency in slower technologies, significant efforts are dedicated to mitigate the noise introduced by upconverters, such as the mixers 135, 155. In contrast, the baseband signal path (for example, the path starting at the data input 105 and ending at the mixers 135, 155) was often a relatively easy part of traditional transmitter design because large signal swings allowed a high signal-to-noise ratio (SNR) with reasonable power dissipation.
As an example, typical amplifier topologies can support a maximum peak-to-peak signal swing of Vdd−3*Vdsat. Typically, the saturation voltage Vdsat is on the order of 200 mV. Comparing the peak baseband signal swing achievable with a 2.7 V supply to that with a 1.2 V supply, the 1.2 V implementation can tolerate 3.5 times less signal swing. Since the thermal noise current of a transistor is proportional to the square-root of the drain current, 3.5 times less signal swing translates to more than 12 times higher current for a given SNR. Such a current is unacceptable for many applications, and design constraints may prevent relaxation of the signal-to-noise ratio.
In addition to the large signal swing that older technologies with higher power supplies made possible, circuit techniques for making linear transconductors can alternatively be used. For example, resistive degeneration can be used to linearize the voltage-to-current transfer function of a MOSFET or bipolar transistor. In the case of a MOSFET, the voltage drop across the degeneration resistor is typically large compared to Vdsat in order for the degeneration to be effective. With the 1.2 V power supply of, for example, a 65 nm CMOS process, resistor degeneration is not practical.
FIG. 2 is a graph 200 depicting the frequency response of a transmitter, such as the transmitter 100 shown in FIG. 1. The graph 200 shows the power output by the transmitter (Pout) along the vertical axis at various frequencies (f) along the horizontal axis. As shown in the graph 200, there is a power output peak 210 of width BW, centered around the frequency of the local oscillator (fLO). BW is the target bandwidth for a given application. Ideally, there is no transmission outside of this band. In practice, however, a transmitter will typically transmit some out-of-band noise 220, which falls off at frequencies above and below fLO. The SNR at frequency offset Δf is defined by the height of the peak 210 divided by the noise at Δf offset from fLO.